Enhanced doherty amplifier

ABSTRACT

The disclosure relates to an enhanced Doherty amplifier that provides significant performance improvements over conventional Doherty amplifiers. The enhanced Doherty amplifier includes a power splitter, combining node, a carrier path, and a peaking path. The power splitter is configured to receive an input signal and split the input signal into a carrier signal provided at a carrier splitter output and a peaking signal provided at a peaking splitter output. The carrier path includes carrier power amplifier circuitry, a carrier input network coupled between the carrier splitter output and the carrier power amplifier circuitry, and a carrier output network coupled between the carrier power amplifier circuitry and the Doherty combining node. The peaking path includes peaking power amplifier circuitry, a peaking input network coupled between the peaking splitter output and the peaking power amplifier circuitry, and a carrier output network coupled between the power amplifier circuitry and the Doherty combining node.

CROSS REFERENCE TO RELATED APPLICATION

The present application is a continuation of U.S. patent applicationSer. No. 13/049,312, which was filed on Mar. 16, 2011, the disclosure ofwhich is incorporated herein by reference in its entirety

FIELD OF THE DISCLOSURE

The present disclosure relates to power amplifiers and in particular toDoherty amplifiers that are capable of operating efficiently over widerbandwidths than conventional Doherty amplifiers.

BACKGROUND

As current mobile communication systems evolve and new communicationssystems are developed, there is continuing demand for more powerful andefficient power amplifiers that are capable of operating over broaderfrequency ranges. Many of these communication systems employ mobiledevices and access points, such as base stations, that are batterypowered. For such communication devices, more efficient power amplifiersyield longer operating times between battery charges.

Further, the transmit power levels for mobile devices and especiallyaccess points are continuing to increase at the same time that sizes ofthese devices are shrinking. As the power levels increase, the amount ofheat that is generated during amplification generally increases.Therefore, designers are faced with dissipating greater quantities ofheat from shrinking communication devices or reducing the amount of heatgenerated by the power amplifiers therein. More efficient poweramplifiers are preferred because they generate less heat than lessefficient power amplifiers at corresponding power levels, and thusreduce the amount of heat to dissipate during operation.

Given the ever increasing demand for efficiency, the Doherty amplifierhas become a popular power amplifier in mobile communicationapplications, especially base station applications. While relativelyefficient compared to its rivals, the Doherty amplifier has a relativelylimited bandwidth of operation. For example, a well-designed Dohertyamplifier may provide an instantaneous bandwidth of 5 percent, whichcorresponds to about 100 MHz for a 2 GHz signal and is generallysufficient to support a single communication band. For example,Universal Mobile Telecommunications Systems (UMTS) devices operate in aband between 2.11 and 2.17 GHz, and thus require an instantaneousbandwidth of 60 MHz (2.17 GHz-2.11 GHz). A Doherty amplifier can beconfigured to support an instantaneous bandwidth of 60 MHz for the UMTSband. Accordingly, for communication devices that only need to support asingle communication band, the limited operating bandwidth of theDoherty power amplifier poses no problems.

However, modern communication devices are often required to supportvarious communication standards that employ different modulationtechniques over a wide range of operating frequencies. These standardsinclude but are not limited to the Global System for MobileCommunications (GSM), Personal Communication Service (PCS), UniversalMobile Telecommunications Systems (UMTS), Worldwide Interoperability forMicrowave Access (WiMAX), Long Term Evolution (LTE), and the like.

The bands of operation for these standards range from around 800 MHz to4 GHz for consumer telecommunication applications and from 20 MHz to 6GHz for military applications. The GSM standards alone employ bandsranging from around 800 MHz to 2 GHz. For example, GSM-850 uses an824-894 MHz band, GSM-900 uses an 890-960 MHz band, GSM-1800 uses a1710-1880 MHz band, and GSM-1900 uses an 1850-1990 MHz band. UMTS uses a2.11-2.17 GHz band. LTE uses a 2.6-2.7 GHz band, and WiMAX uses bandscentered about 2.3, 2.5, 3.3 and 3.5 GHz. Thus, for devices that need tosupport multiple communication bands, a single Doherty amplifier is notsufficient.

For communication devices that support multiple standards over disparatecommunication bands, designers often employ multiple power amplifierchains for each of the different communication bands, which increasesthe size, cost, and complexity of the communication devices. As such,there is a need to increase the effective operating range of a Dohertypower amplifier to support multiple communication bands, which arespread over a significant frequency range, while maintaining theefficiency afforded by current Doherty power amplifier designs.

SUMMARY

The present disclosure relates to an enhanced Doherty amplifier thatprovides significant performance improvements over conventional Dohertyamplifiers. The enhanced Doherty amplifier includes a power splitter, acombining node, a carrier path, and a peaking path. The power splitteris configured to receive an input signal and split the input signal intoa carrier signal provided at a carrier splitter output and a peakingsignal provided at a peaking splitter output. The carrier path includescarrier power amplifier circuitry, a carrier input network coupledbetween the carrier splitter output and the carrier power amplifiercircuitry, and a carrier output network coupled between the carrierpower amplifier circuitry and the Doherty combining node. The peakingpath includes peaking power amplifier circuitry, a peaking input networkcoupled between the peaking splitter output and the peaking poweramplifier circuitry, and a carrier output network coupled between thepower amplifier circuitry and the Doherty combining node.

In one embodiment, the carrier and peaking input networks are configuredto impose phase shifts causing the peaking signal to lag the carriersignal by approximately 90 degrees when the carrier and peaking signalsare respectively presented to the carrier and peaking power amplifiercircuitries. The carrier and peaking output networks are configured toimpose further phase shifts causing the peaking and carrier signals toarrive at the Doherty combining node for reactive combining to generatean output signal. The carrier input and output networks and the peakinginput and output networks may include lumped elements and need notinclude transmission lines. As such, these networks may be synthesizedas a group to provide improved performance characteristics for theoverall enhanced Doherty amplifier.

Those skilled in the art will appreciate the scope of the disclosure andrealize additional aspects thereof after reading the following detaileddescription in association with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings incorporated in and forming a part of thisspecification illustrate several aspects of the disclosure, and togetherwith the description serve to explain the principles of the disclosure.

FIG. 1 is schematic diagram of a conventional Doherty amplifier.

FIG. 2 is a plot of input power versus output power for the carrier andpeaking amplifier circuitries of the conventional Doherty amplifier.

FIG. 3A is a plot of efficiency versus output power for a typical(non-Doherty) power amplifier.

FIG. 3B is a plot of efficiency versus output power for a conventionalDoherty amplifier.

FIG. 4A is a plot of gain versus frequency for a wideband (non-Doherty)power amplifier.

FIG. 4B is a plot of gain versus frequency for a conventional Dohertyamplifier employing wideband amplifiers.

FIG. 5 is schematic diagram of an enhanced Doherty amplifier, accordingto one embodiment of the disclosure.

FIG. 6A is a plot of efficiency versus frequency for a firstconfiguration of the enhanced Doherty amplifier of FIG. 5.

FIG. 6B is a plot of peak output power versus frequency for the firstconfiguration of the enhanced Doherty amplifier of FIG. 5.

FIG. 7A is a plot of efficiency versus frequency for a secondconfiguration of the enhanced Doherty amplifier of FIG. 5.

FIG. 7B is a plot of peak output power versus frequency for the secondconfiguration of the enhanced Doherty amplifier of FIG. 5.

FIG. 8 is schematic diagram of an enhanced Doherty amplifier, accordingto another embodiment of the disclosure.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the disclosure andillustrate the best mode of practicing the disclosure. Upon reading thefollowing description in light of the accompanying drawings, thoseskilled in the art will understand the concepts of the disclosure andwill recognize applications of these concepts not particularly addressedherein. It should be understood that these concepts and applicationsfall within the scope of the disclosure and the accompanying claims.

The present disclosure relates to increasing the bandwidth of operationof a Doherty power amplifier. Prior to delving into the details of how aDoherty power amplifier can be modified to increase its bandwidth ofoperation, an overview of a traditional Doherty power amplifier 10 isprovided in association with FIG. 1. As illustrated, a modulated RFinput signal RF_(IN) is fed to a power splitter 12, such as a Wilkinsonsplitter, which splits the RF input signal RF_(IN) along a “carrierpath” and a “peaking path.” Traditionally, the RF input signal RF_(IN)is split evenly such that the carrier path and the peaking path receiveone half (−3 dB) of the original input power of the RF input signalRF_(IN).

The carrier path generally includes carrier power amplifier circuitry(PA_(C)) 14 followed by a first transmission line (TL) 16 that is sizedto provide a 90° phase shift at or near the center frequency of theoperating bandwidth. The carrier path terminates at a Doherty combiningnode 18, which is further coupled to a transformer 24, which isultimately coupled to an antenna (not shown). The peaking path includesa second transmission line (TL) 20 that is sized to provide a 90° phaseshift at or near the center frequency of the operating bandwidthfollowed by peaking power amplifier circuitry (PA_(P)) 22. As such, theRF input signal RF_(IN) provided along both the carrier path and thepeaking path are 90° out of phase with one another when they areamplified by the respective carrier and peaking power amplifiercircuitries 14 and 22. As with the carrier path, the peaking pathterminates into the Doherty combining node 18. Notably, the powersplitter 12 may inherently provide a 90° phase shift in the leg feedingthe peaking path. In such cases, the second transmission line 20 is notincluded.

In traditional Doherty fashion, the carrier power amplifier circuitry 14provides a class NB (or B) amplifier, and the peaking power amplifiercircuitry 22 provides a class C amplifier. During operation, the RFinput signal RF_(IN) is split and directed along the carrier and peakingpaths to the respective carrier and peaking power amplifier circuitries14 and 22. Notably, the second transmission line 20 delays the portionof the RF input signal RF_(IN) in the peaking path by 90° prior toreaching the peaking power amplifier circuitry 22.

A Doherty amplifier is generally considered to have two regions ofoperation. In the first region, only the carrier power amplifiercircuitry 14 is turned on and operates to amplify the RF input signalRF_(IN). In the second region, both the carrier power amplifiercircuitry 14 and the peaking power amplifier circuitry 22 operate toamplify the RF input signal RF_(IN) in the respective carrier andpeaking paths. The threshold between the two regions corresponds to amagnitude of RF input signal RF_(IN) in the carrier path where carrierpower amplifier circuitry 14 becomes saturated. In the first region, thelevels of the RF input signal RF_(IN) are below the threshold. In thesecond region, the levels of RF input signal RF_(IN) are at or above thethreshold

In the first region where the level of the RF input signal RF_(IN) isbelow the given threshold, the carrier power amplifier circuitry 14amplifies the portion of the RF input signal RF_(IN) in the carrierpath. When the RF input signal RF_(IN) is below the given threshold, thepeaking power amplifier circuitry 22 is turned off and consumes littlepower. As such, only the carrier power amplifier circuitry 14 suppliesan amplified RF input signal RF_(IN) to the Doherty combining node 18and transformer 24 to provide an RF output signal RF_(OUT). The overallefficiency of the Doherty amplifier is determined predominantly by theefficiency of the class AB (or B) amplifier of the carrier poweramplifier circuitry 14.

In the second region where the RF input signal RF_(IN) is at or abovethe given threshold, the carrier power amplifier circuitry 14 issaturated and delivers its maximum power to the Doherty combining node18 via the first transmission line 16. Further, as the RF input signalRF_(IN) rises above the given threshold, the peaking power amplifiercircuitry 22 turns on and begins amplifying the portion of the RF inputsignal RF_(IN) that flows along the peaking path. As the RF input signalRF_(IN) continues to rise above the given threshold, the peaking poweramplifier circuitry 22 delivers more power to the Doherty combining node18 until the peaking power amplifier circuitry 22 becomes saturated.

In the second region, both the carrier and peaking power amplifiercircuitries 14 and 22 are delivering amplified signals to the Dohertycombining node 18. By employing the first and second transmission lines16 and 20 in the carrier and peaking paths, the amplified signals ineach path reach the Doherty combining node in phase and are reactivelycombined. The combined signal is then stepped up or down via thetransformer 24 to generate the amplified RF output signal RF_(OUT).

The graph of FIG. 2 plots output power (P_(O)) versus input power(P_(I)) for the carrier power amplifier circuitry 14, the peaking poweramplifier circuitry 22, and the overall Doherty amplifier 10. Asillustrated, the carrier power amplifier circuitry 14 operates linearlythroughout first region R1 until becoming saturated. Once the carrierpower amplifier circuitry 14 reaches saturation, the second region R2 isentered. In the second region R2, the peaking power amplifier circuitry22 turns on and begins to amplify the RF input signal RF_(IN). Theoverall output power for the Doherty amplifier is effectively the sum ofthe output power of the carrier and peaking power amplifiers 14 and 22in the second region R2.

When operating in the second region R2, the power supplied by thepeaking power amplifier circuitry 22 effectively reduces the apparentload impedance presented to the carrier power amplifier circuitry 14.Reducing the apparent load impedance allows the carrier power amplifiercircuitry 14 to deliver more power to the load while remainingsaturated. As a result, the maximum efficiency of the carrier poweramplifier circuitry 14 is maintained and the overall efficiency of theDoherty amplifier 10 remains high throughout the second region R2 untilthe peaking power amplifier circuitry 22 becomes saturated.

The graphs of FIGS. 3A and 3B plot efficiency versus output power for atypical power amplifier and a typical Doherty amplifier, respectively.With reference to FIG. 3A, the efficiency η of the typical poweramplifier increases proportionally to the output power P until the poweramplifier saturates and reaches its maximum output power P_(MAX). Asillustrated in FIG. 3B, the carrier power amplifier circuitry 14 of theDoherty amplifier 10 operates in a similar fashion. Progressing throughfirst region R1, the peaking power amplifier circuitry 22 remains offand the RF input signal RF_(IN) increases to a point where the carrierpower amplifier circuitry 14 becomes saturated. Throughout the firstregion R1, the efficiency of the carrier power amplifier circuitry 14,and thus the overall efficiency η for the Doherty amplifier 10,increases proportionally with the output power P until the carrier poweramplifier circuitry 14 becomes saturated at a given output power level.This given output power level is referred to herein as a threshold powerlevel P_(TH), and is shown, for illustrative purposes only, at one-ninth( 1/9) of the maximum output power P_(MAX) ( 1/9 P_(MAX)) of the Dohertyamplifier 10.

As the RF input signal RF_(IN) increases past the point where thecarrier power amplifier circuitry 14 becomes saturated, the Dohertyamplifier enters the second region R2. As the second region R2 isentered, the peaking power amplifier circuitry 22 begins to amplify theRF input signal RF_(IN). The carrier power amplifier circuitry 14remains saturated and continues to amplify the RF input signal RF_(IN).As the RF input signal RF_(IN) increases further, the peaking poweramplifier circuitry 22 delivers more power until the peaking poweramplifier circuitry 22 becomes saturated at the maximum output powerP_(max) of the Doherty amplifier 10. Throughout the second region R2,the overall efficiency η for the Doherty amplifier 10 remains high andpeaks at the beginning of the second region R2 where the carrier poweramplifier circuitry 14 first becomes saturated and at the end of thesecond region R2 where the peaking power amplifier circuitry 22 becomessaturated. As clearly depicted in FIGS. 3A and 3B, the power addedefficiency at backed off power levels from around the threshold powerlevel P_(TH) up to the maximum output power P_(MAX) is significantlyimproved in the Doherty amplifier 10 over that of a typical poweramplifier.

Returning to FIG. 1, the illustrated Doherty amplifier 10 is shown tohave a third transmission line 26 in the carrier path and a fourthtransmission line 28 in the peaking path. The third and fourthtransmission lines 26 and 28 may be used to provide phase offsets in theoutputs of the carrier and peaker paths in an effort to have thechanging output impedance of the peaking power amplifier circuitry 22properly load the output impedance of the carrier power amplifiercircuitry 14 and vice versa.

As shown above, a conventional Doherty amplifier 10 is very efficient atboth heavily backed off and maximum power levels. Unfortunately, theconventional Doherty amplifier 10 is relatively bandwidth limited andonly provides an available instantaneous bandwidth of 5% of theoperating frequency. For example, a Doherty amplifier 10 designed totransmit signals centered around 2.1 GHz will have at most an availablebandwidth of approximately 105 MHz.

Notably, the carrier and peaking power amplifier circuitries 14 and 22do not limit the bandwidth of the conventional Doherty amplifier 10.Even if these carrier and peaking power amplifier circuitries 14 and 22were designed to be wideband amplifiers and individually supportbandwidths of several octaves, the overall instantaneous bandwidth ofthe conventional Doherty amplifier 10 would remain limited to around 5%of the operating frequency. For example, if each of the carrier andpeaking power amplifier circuitries 14 and 22 were individually designedto have available bandwidths between 2 GHz to 4 GHz, the overallinstantaneous bandwidth of the Doherty amplifier 10 would remain limitedto around 5% of the operating frequency (100 MHz at 2 GHz; 400 MHz at 6Hz). Thus, no matter how wide the operating range of the carrier andpeaking power amplifier circuitries 14 and 22 that you employ in theconventional Doherty amplifier 10, other components of the conventionalDoherty amplifier 10 limit the available bandwidth.

FIGS. 4A and 4B illustrate the above concept. FIG. 4A plots gain versusfrequency for a wideband power amplifier, and FIG. 4B plots gain versusfrequency for the conventional Doherty amplifier 10 where the samewideband power amplifier is used for both the carrier and peaking poweramplifier circuitries 14 and 22. As depicted, the conventional Dohertyamplifier 10 has a much more limited bandwidth than that of thestand-alone wideband power amplifier, even when it employs widebandamplifiers in the carrier and peaking power amplifier circuitries 14 and22. Thus, simply using a wideband power amplifier in the conventionalDoherty amplifier 10 will not necessarily increase the bandwidth of theDoherty amplifier 10.

It has been discovered that the primary bandwidth limiting components ofthe conventional Doherty amplifier 10 are the power splitter 12, thefirst and second transmission lines 16, 20 that provide the 90° phaseshifts, the third and fourth transmission lines 26, 28 that provide thephase offsets, and the transformer 24. The present disclosure providestechniques for replacing or modifying various components of theconventional Doherty amplifier 10 to significantly increase the overallbandwidth of the conventional Doherty amplifier 10.

An example of an enhanced Doherty amplifier 30 is illustrated in FIG. 5.In particular, a modulated RF input signal RF_(IN) is fed to a powersplitter 32, such as a Wilkinson splitter, which splits the RF inputsignal RF_(IN) along the carrier path and the peaking path. In thisexample, the RF input signal RF_(IN) is unevenly split such that thecarrier path receives the input power of the RF input signal RF_(IN)attenuated by 1.7 dB and the peaking path receives the input power ofthe RF input signal RF_(IN) attenuated by 4.7 dB. An uneven split inthis manner further increases the efficiency of the enhanced Dohertyamplifier 30 relative to an even split wherein with an even split, theRF input signal RF_(IN) is split evenly (−3 dB) between the carrier andpeaking paths.

The carrier path includes a carrier input network 34, carrier poweramplifier circuitry (PA_(C)) 36, and a carrier output network 38. Thecarrier path terminates at a Doherty combining node 40, which is furthercoupled to a transformer 42, which is ultimately coupled to an antenna(not shown). The peaking path includes a peaking input network 44,peaking power amplifier circuitry (PA_(C)) 46, and a peaking outputnetwork 48. The peaking path terminates at the Doherty combining node40.

In this example, the split RF input signals RF_(IN) that are provided bythe power splitter 32 are presented to the carrier and peaking inputnetworks 34, 44 substantially in phase. In other words, the powersplitter does not impart a 90° phase shift to the RF input signalRF_(IN) that is provided to the peaking path in this embodiment.However, the RF input signals RF_(IN) that are provided to therespective inputs of the carrier and peaking power amplifier circuitries36, 46 need to by shifted by approximately 90°. Generally, the RF inputsignal RF_(IN) that is presented to the input of the peaking poweramplifier circuitry 46 lags the RF input signal RF_(IN) that ispresented to the input of the carrier power amplifier circuitry 36 byapproximately 90°.

In one embodiment, the carrier and peaking input networks 34, 44 arelumped element networks that are designed to ensure that the RF inputsignal RF_(IN) that is presented to the input of the peaking poweramplifier circuitry 46 lags the RF input signal RF_(IN) that ispresented to the input of the carrier power amplifier circuitry 36 byapproximately 90°. A lumped element network is one that includesinductors, capacitors, and resistors as the primary filtering and phaseshifting components. In the illustrated embodiment, the carrier inputnetwork 34 advances the RF input signal RF_(IN) in the carrier path by45°) (+45°, and the peaking input network 44 delays the RF input signalRF_(IN) in the peaking path by 45°) (−45°. By advancing the RF inputsignal RF_(IN) in the carrier path by 45° and delaying the RF inputsignal RF_(IN) in the peaking path by 45° (−45°), the RF input signalRF_(IN) that is presented to the input of the peaking power amplifiercircuitry 46 lags the RF input signal RF_(IN) that is presented to theinput of the carrier power amplifier circuitry 36 by approximately 90°.

While phase shifts of +45° and −45° in the respective carrier andpeaking input networks 34, 44, are described, other combinations ofphase shifts are possible. For example, phase shifts of +60° and −30° or−50° and +40° in the respective carrier and peaking input networks 34,44 may be employed.

The carrier input network 34 of FIG. 5 is shown to include a seriescapacitor C₁, a shunt inductor L₁, and a series capacitor C₂. Thepeaking input network 44 is shown to include a series inductor L₂, ashunt capacitor C₃, and a series inductor L₃. As will be appreciated byone skilled in the art, these networks are merely exemplary and may beimplemented in higher order (second and third order) networks of variousconfigurations.

Continuing with FIG. 5, the carrier output network 38 is coupled betweenthe carrier power amplifier circuitry 36 and the Doherty combining node40. Similarly, the peaking output network 48 is coupled between thepeaking power amplifier circuitry 46 and the Doherty combining node 40.The primary functions of the carrier and peaking output networks 38, 48are to remove the phase shifts provided by the carrier and peaking inputnetworks 34, 44 and provide any phase offsets deemed necessary toachieve desired performance metrics. After passing through the carrierand peaking output networks 38, 48, the amplified RF input signalsRF_(IN) from the carrier and peaking paths are presented to the Dohertycombining node 40 in a phase alignment that allows the signals to beefficiently combined and stepped up or down by the transformer 42. Afteramplification, the RF input signal RF_(IN) presented to the peakingoutput network 48 lags the RF input signal RF_(IN) presented to thecarrier output network 38 by approximately 90°. In the illustratedembodiment, the carrier output network 38 effectively shifts the RFinput signal RF_(IN) in the carrier path by a compensated carrier phaseshift φ_(C-COMP).

The compensated carrier phase shift φ_(C-COMP) is the negative of thephase shift provided by the carrier input network 34 (φ_(C-IP)) minus acarrier phase offset φ_(C-PO), wherein φ_(C-COMP)=φ_(C-IP)−φ_(C-PO). Inthis example, the phase shift provided by the carrier input network 34(φ_(C-IP)) is +45°. The carrier phase offset φ_(C-PO) corresponds to thereactive component of the impedance presented to the output of thecarrier power amplifier circuitry 36 at the intended operating frequencyrange or ranges. This impedance is effectively the composite impedanceprovided by the carrier output network 38, the peaking path, and thetransformer 42 at the intended operating frequency range or ranges. Thegoal is to have a substantially real impedance (pure resistive)presented to the output of the carrier power amplifier circuitry 36 atthe intended operating frequency range or ranges.

Similarly, the peaking output network 48 effectively shifts the RF inputsignal RF_(IN) in the peaking path by a compensated peaking phase shiftφ_(P-COMP). The compensated peaking phase shift φ_(P-COMP) is thenegative of the phase shift provided by the peaking input network 44(φ_(P-IP)) minus a peaking phase offset φ_(P-PO), whereinφ_(P-COMP)=−φ_(P-IP)−φ_(P-PO). In this example, the phase shift providedby the peaking input network 44 (φ_(P-IP)) is −45°. The peaking phaseoffset φ_(P-PO) corresponds to the reactive component of the impedancepresented to the output of the peaking power amplifier circuitry 46 atthe intended operating frequency range or ranges. This impedance iseffectively the composite impedance provided by the peaking outputnetwork 48, the carrier path, and the transformer 42 at the intendedoperating frequency range or ranges. The goal is to have a substantiallyreal impedance (pure resistive) presented to the output of the peakingpower amplifier circuitry 46 at the intended operating frequency rangeor ranges. While baseline phase shifts of +45° (φ_(C-IP)) and −45°(φ_(P-IP)) in the respective carrier and peaking output networks 38, 48,are described, these phase shifts merely mirror those provided in therespective carrier and peaking input networks 34, 44. As noted above,other combinations of phase shifts are possible.

In FIG. 5, the carrier output network 38 is shown to include a seriesinductor L₄, a shunt capacitor C₄, and a series inductor L₅. The peakingoutput network 48 is shown to include a series capacitor C₅, a shuntinductor 4, and a series capacitor C₆. As will be appreciated by oneskilled in the art, these networks are merely exemplary and may beimplemented in higher order networks of various configurations.

In the illustrated embodiment, the carrier power amplifier circuitry 36provides a class A/B (or B) amplifier, and the peaking power amplifiercircuitry 46 provides a class C amplifier. Each of these amplifiers isgenerally formed from one or more transistors. In select embodiments,the amplifiers are formed from one of Gallium Nitride (GaN) highelectron mobility transistors (HEMTs), Gallium Arsenide (GaAs) orSilicon Carbide (SiC) metal semiconductor field effect transistor(MESFETS), and laterally diffused metal oxide semiconductor (LDMOS)transistors. However, those skilled in the art will recognize otherapplicable transistors and material systems are applicable.

During operation of the enhanced Doherty amplifier 30, the RF inputsignal RF_(IN) is split by the power splitter 32 and directed along thecarrier and peaking paths to the respective carrier and peaking poweramplifier circuitries 36 and 46. The RF input signal RF_(IN) is advanced45° in the carrier path by the carrier input network 34 before beingpresented to the carrier power amplifier circuitry 36. The RF inputsignal RF_(IN) is delayed 45° in the peaking path by the peaking inputnetwork 48 before being presented to the peaking power amplifiercircuitry 46.

As noted above, Doherty amplifiers characteristically operate in tworegions. In the first region R1, only the carrier power amplifiercircuitry 36 is turned on and operates to amplify the RF input signalRF_(IN). In the second region R2, both the carrier power amplifiercircuitry 36 and the peaking power amplifier circuitry 46 operate toamplify the RF input signal RF_(IN) in the respective carrier andpeaking paths. The threshold between the two regions corresponds to amagnitude of RF input signal RF_(IN) in the carrier path where thecarrier power amplifier circuitry 36 becomes saturated. In the firstregion R1, the levels of the RF input signal RF_(IN) are below thethreshold. In the second region R2, the levels of RF input signalRF_(IN) are at or above the threshold

In the first region R1 where the level of the RF input signal RF_(IN) isbelow the given threshold, the carrier power amplifier circuitry 36amplifies the portion of the RF input signal RF_(IN) in the carrierpath. The amplified RF input signal RF_(IN) is shifted by thecompensated carrier phase shift φ_(C-COMP) by the carrier output network38 and passed to the Doherty combining node 40. Notably, effectively nosignal is provided to the Doherty combing 40 node via the peaking pathin the first region R1 when the RF input signal RF_(IN) is below thegiven threshold. In the first region R1, the peaking power amplifiercircuitry 46 is turned off and the overall efficiency of the enhancedDoherty amplifier 30 is determined predominantly by the efficiency ofthe carrier power amplifier circuitry 36.

In the second region R2 where the RF input signal RF_(IN) is at or abovethe given threshold, the carrier power amplifier circuitry 36 issaturated and delivers its maximum power to the Doherty combining node40 via the carrier output network 38. Again, the amplified RF inputsignal RF_(IN) is shifted by the compensated carrier phase shiftφ_(C-COMP) by the carrier output network 38 and passed to the Dohertycombining node 40.

Further, as the RF input signal RF_(IN) rises above the given threshold,the peaking power amplifier circuitry 46 turns on and begins amplifyingthe portion of the RF input signal RF_(IN) that flows along the peakingpath. As the RF input signal RF_(IN) continues to rise above the giventhreshold, the peaking power amplifier circuitry 46 delivers more powerto the Doherty combining node 40 via the peaking output network 48 untilthe peaking power amplifier circuitry 46 becomes saturated. Notably, thepeaking output network 48 effectively shifts the RF input signal RF_(IN)in the peaking path by the compensated peaking phase shift φ_(P-COMP).Accordingly, the RF input signals RF_(IN) arrive at the Dohertycombining node 40 from the respective carrier and peaking paths, arereactively combined at the Doherty combining node 40, and are thenstepped up or down via the transformer 42 to generate the RF outputsignal RF_(OUT).

In comparison with the conventional Doherty amplifier 10 (FIG. 1), theenhanced Doherty amplifier 30 (FIG. 5) has effectively replaced thetransmission lines 16, 20, 26, 28 with the input and output networks 34,44, 38, 48 in both the carrier and peaking paths. Employing lumpedelement-based input and output networks 34, 44, 38, 48 in the carrierand peaking paths allows the enhanced Doherty amplifier 30 to be viewedand synthesized as a band-pass filter. As such, the respective networksas well as the power splitter 32 and transformer 42 may be synthesizedas part of the enhanced Doherty amplifier 30 to achieve desiredperformance characteristics in much the same fashion as a band-passfilter can be synthesized. The performance characteristics of primaryinterest in the enhanced Doherty amplifier 30 include bandwidth,terminal impedances, power gain, and output power.

While the input and output networks 34, 44, 38, 48 may be synthesized toemulate the amplitude and phase responses of the transmission lines 16,20, 26, 28, doing so would limit the performance of the enhanced Dohertyamplifier 30 to that of the conventional Doherty amplifier 10. Forenhanced performance, the order and configuration of the input andoutput networks 34, 44, 38, 48 may be synthesized to better optimize thephase differences between the carrier and peaking paths as well asprovide improved input and output matching to achieve desiredperformance characteristics at maximum and backed-off power levels.Notably, the effective bandwidth of the enhanced Doherty amplifier 30can be dramatically increased over what has been achieved by theconventional Doherty amplifier 10 while maintaining high efficiency atmaximum and backed-off power levels.

This increase in bandwidth can be used to allow a single enhancedDoherty amplifier 30 to cover multiple communication bands that operatein disparate frequency bands, increase the available bandwidth for agiven communication band to support higher data rates and additionalchannels, or a combination thereof. As noted above, the conventionalDoherty amplifier 10 is relatively bandwidth limited and only providesan available instantaneous bandwidth of 5% of the operating frequency.For example, UMTS is allocated the frequency band of 2.11 and 2.17 GHzand requires a minimum bandwidth of 60 MHz. Since the conventionalDoherty amplifier 10 can support a bandwidth of 105 MHz, it can handlethe UMTS band. However, if there is a need to handle the UMTS bandbetween 2.11 and 2.17 GHz as well as LTE band between 2.6 and 2.7 withthe same amplifier circuitry, a bandwidth of essentially 600 MHz isrequired, and clearly, the conventional Doherty amplifier 10 is unableto meet such bandwidth requirements. The enhanced Doherty amplifier 30can be designed to meet these requirements while achieving desirableefficiency, gain, and output power requirements.

The following provides two of many examples where the enhanced Dohertyamplifier 30 can be configured to handle both the UMTS and LTE bands,which reside in the 2.11 to 2.17 GHz and 2.6 to 2.7 GHz bands. For thefirst example, the enhanced Doherty amplifier 30 is synthesized toprovide relatively uniform gain and backed-off power efficiencythroughout a 600 MHz band between 2.11 and 2.7 GHz to cover both theUMTS and LTE bands. As illustrated in FIG. 6A, which is a plot ofefficiency versus frequency at a 6 dB backed-off power level, theenhanced Doherty amplifier 30 can be synthesized to provide relativelyuniform efficiency throughout the 2.11 to 2.7 GHz frequency range atbacked-off power levels. However, since the ultimate bandwidth potentialfor an amplifier design depends on the competing characteristics ofefficiency, gain, and output power, compromises among thesecharacteristics always come into play. In this example, the compromisesresult in a noticeable, but acceptable, drop in peak output power in theLTE band (2.6-2.7 GHz) relative to the peak output power in the UMTSband. The drop is illustrated in FIG. 6B, which plots peak output powerversus frequency.

For the second example, again assume that there is a need to supportboth the UMTS and LTE bands; however, additional output power in the LTEband and higher efficiency is desired when operating in both the UMTSand LTE bands. Further, assume that the efficiency, gain, and outputpower between the UMTS and LTE bands is either unimportant or that thereis a desire to intentionally reduce the gain between the UMTS and LTEbands (≅2.12 to 2.5 GHz). By properly synthesizing the input and outputnetworks 34, 44, 38, 48 and potentially the power splitter 32 andtransformer 42, a tailored response may be achieved. As illustrated inFIG. 7A, which is a plot of efficiency versus frequency at a 6 dBbacked-off power level, the enhanced Doherty amplifier 30 can besynthesized to provide an efficiency response that is optimized for theUMTS and LTE bands. As such, efficiency peaks about the UMTS and LTEbands and dips significantly in the unused frequency band between theUMTS and LTE bands.

Similarly, the peak output power response as of function of frequencyfor the UMTS and LTE bands is also optimized, as illustrated in FIG. 7B.As with efficiency, the peak output powers in the UMTS and LTE bands areboosted relative to the first example (FIG. 6B) while a dip, or null, inthe peak output power (and likely gain) is provided between the UMTS andLTE bands. The dip may also be tailored to help reduce noise orinterference between the bands. In essence, the enhanced Dohertyamplifier 30 can be tailored to trade uniform power and efficiencyacross a wide bandwidth for exceptional frequency and peak output powerresponses in select pass-bands that are separated by wide frequencyrange.

While UMTS and LTE bands are illustrated, other communication bands forthe various standards may be addressed in similar fashion. For example,the first communication band could be one of a PCS band, a UMTS band,and a GSM band and the second communication band could be one of an LTEband and a WiMax band. Further, these concepts may be applied fordifferent communication bands in the same standard. For example, oneenhanced Doherty amplifier 30 could be used to support both the 2.5 and3.5 GHz WiMax bands. Also, a given pass-band may be widened to supportrelatively adjacent communication bands, such as 1.8 GHz PCS and 2.1 GHzUMTS. While only two communication bands are illustrate in the secondexample, the enhanced Doherty amplifier 30 could be synthesized tosupport three or more bands in similar fashion wherein dips inbacked-off power efficiency, gain, or output power may be provided if,and as, desired.

The input and output networks 34, 44, 38, 48 may also be synthesized toprovide responses that have different efficiency, gain, or output powerresponses for different communication bands. For example, forcommunication bands that are separated by 200 MHz, 250 MHz, 300 MHz, 400MHz, 500 MHz, 1 GHz or more, an enhanced Doherty amplifier 30 couldsupport a lower band that is 150 MHz wide at a higher backed-off powerefficiency and peak output power and a higher band that is 250 MHz wideat a slightly lower backed-off power efficiency and peak output power.In essence, the enhanced Doherty amplifier 30 allows for highlyconfigurable responses while providing exceptional efficiency atbacked-off and maximum power levels throughout wide frequency ranges aswell as for disparate communication bands that are separated by largefrequency ranges. Thus, a single power amplifier topology can be used toefficiency support multiple, disparate communication bands.

The enhanced Doherty amplifier 30 is modular, and as such, can be usedin parallel with one or more other enhanced Doherty amplifiers 30 forhigher power applications. An exemplary modular Doherty configuration 50is illustrated in FIG. 8. With the modular Doherty configuration 50, thesame benefits and configurability as described above apply. The modularDoherty configuration 50 includes two enhanced Doherty modules 52A, 52B,which correspond to the enhanced Doherty amplifier 30 of FIG. 5.

An RF input signal RF_(IN) is fed to a power splitter 54, such as aWilkinson splitter, which splits the RF input signal RF_(IN) along twopaths. The first path leads to the input of a power splitter 32A, andthe second path lead to the input of a power splitter 32B. In thisembodiment, the RF input signal RF_(IN) is evenly split between the twopaths such that each of the enhanced Doherty modules 52A, 52B receivesthe input power of the RF input signal RF_(IN) attenuated by 3 dB viathe respective power splitters 32A, 32B.

The power splitters 32A, 32B split the RF input signal RF_(IN) along therespective carrier and peaking paths of the enhanced Doherty modules52A, 52B. In this example, the RF input signal RF_(IN) is unevenly splitby the power splitters 32A, 32B, such that the carrier paths receive theinput power of the RF input signal RF_(IN) attenuated by another 1.7 dBand the peaking path receive the input power of the RF input signalRF_(IN) attenuated by another 4.7 dB. As noted above, employing anuneven split in this manner further increases the efficiency of theenhanced Doherty amplifier relative to an even split.

The carrier paths of the respective enhanced Doherty modules 52A, 52Binclude carrier input networks 34A, 34B, carrier power amplifiercircuitries (PA_(C)) 36A, 36B, and carrier output networks 38A, 38B. Thecarrier paths terminate at respective Doherty combining nodes 40A, 40B,which are further coupled to respective transformers 42A, 42B. Thepeaking paths include peaking input networks 44A, 44B, peaking poweramplifier circuitries (PA_(P)) 46A, 46B, and peaking output networks48A, 48B. The peaking paths terminate at the respective Dohertycombining nodes 40A, 40B.

The split RF input signals RF_(IN) that are provided by the powersplitters 32A, 32B are presented to the carrier and peaking inputnetworks 34A, 34B, 44A, 44B substantially in phase. In one embodiment,the carrier and peaking input networks 34A, 34B, 44A, 44B are lumpedelement networks that are designed to ensure that the RF input signalsRF_(IN) that are presented to the input of the peaking power amplifiercircuitries 46A, 46B lag the RF input signals RF_(IN) that are presentedto the input of the carrier power amplifier circuitries 36A, 36B byapproximately 90°. In the illustrated embodiment, the carrier inputnetworks 34A, 34B advance the RF input signals RF_(IN) in the carrierpaths by 45°) (+45°, and the peaking input networks 44A, 44B delay theRF input signals RF_(IN) in the peaking paths by 45°) (−45°. Byadvancing the RF input signals RF_(IN) in the carrier path by 45° anddelaying the RF input signals RF_(IN) in the peaking paths by 45°(−45°), the RF input signals RF_(IN) that are presented to the inputs ofthe peaking power amplifier circuitries 46A, 46B lag the RF inputsignals RF_(IN) that are presented to the inputs of the carrier poweramplifier circuitries 36A, 36B by approximately 90°. While phase shiftsof +45° and −45° in the respective carrier and peaking input networks34A, 34B, 44A, 44B, are described, other combinations of phase shiftsare possible.

The carrier output networks 38A, 38B are coupled between the carrierpower amplifier circuitries 36A, 36B and the respective Dohertycombining nodes 40A, 40B. Similarly, the peaking output networks 48A,48B are coupled between the peaking power amplifier circuitries 46A, 46Band the respective Doherty combining node 40A, 40B. The primaryfunctions of the carrier and peaking output networks 38A, 38B, 48A, 48Bare to remove the phase shifts provided by the carrier and peaking inputnetworks 34A, 34B, 44A, 44B and provide any phase offsets deemednecessary to achieve desired performance metrics. After passing throughthe carrier and peaking output networks 38A, 38B, 48A, 48B, theamplified RF input signals RF_(IN) from the carrier and peaking pathsare presented to the respective Doherty combining nodes 40A, 40B in aphase alignment that allows the signals to be efficiently combined andstepped up or down by the respective transformers 42A, 42B.

After amplification, the RF input signals RF_(IN) presented to thepeaking output networks 48A, 48B lag the RF input signals RF_(IN)presented to the carrier output networks 38A, 38B by approximately 90°.In the illustrated embodiment, the carrier output networks 38A, 38Beffectively shift the RF input signals RF_(IN) in the carrier path by acompensated carrier phase shift φ_(C-COMP). Similarly, the peakingoutput networks 48A, 48B effectively shift the RF input signal RF_(IN)in the peaking path by the compensated peaking phase shift φ_(P-COMP).The carrier and peaking input networks 34A, 34B, 44A, 44B may be second,third, or higher order networks.

Once the signals from the respective carrier and peaking paths arecombined at the Doherty combing nodes 40A, 40B and stepped up or down bythe respective transformers 42A, 42B, the resultant signals from each ofthe enhanced Doherty modules 52A, 52B are combined via the coupler 56 tocreate the RF output signal RF_(OUT).

Each of the Doherty modules 52A, 52B operates in two regions, asdescribed above for the enhanced Doherty amplifier 30. In the firstregion, only the carrier power amplifier circuitries 36A, 36B are turnedon and operate to amplify the RF input signal RF_(IN). In the secondregion, the carrier power amplifier circuitries 36A, 36B and the peakingpower amplifier circuitries 46A, 46B operate to amplify the RF inputsignal RF_(IN) in the respective carrier and peaking paths. Thethreshold between the two regions corresponds to a magnitude of RF inputsignal RF_(IN) in the carrier path where the carrier power amplifiercircuitries 36A, 36B become saturated. In the first region, the levelsof the RF input signal RF_(IN) are below the threshold. In the secondregion, the levels of RF input signal RF_(IN) are at or above thethreshold

As seen from above, the enhanced Doherty amplifiers (30, 50) of thepresent disclosure provide significant performance improvements overconventional Doherty amplifiers designs. Further, the configurability ofthe enhanced Doherty amplifiers (30, 50) allows support for multiplecommunication bands that fall in relatively disparate frequency ranges.These bandwidth improvements are due to the ability to better optimizeimpedance tracking between the carrier and peaking paths as well asimprove the input and output matching relative to the amplifiers in thecarrier and peaking paths. Further, the large signal input and outputreturn losses, both at heavily backed-off and maximum power levels, canbe significantly improved over conventional designs.

While innumerable performance configurations are possible, the followingillustrates some exemplary configurations wherein the enhanced Dohertyamplifier (30, 50) is configured to provide at any one of theaforementioned communication bands:

-   -   an instantaneous bandwidth of at least 15 percent and an        efficiency of greater than 45 percent between 6 dB backed-off        power and peak maximum output power when amplifying radio        frequency signals in either of two different communication bands        employing the same or different communication standards (i.e.        when the communication bands are separated by 300 MHz);    -   an instantaneous bandwidth of at least 15 percent and an        efficiency of greater than 40 percent between 6 dB backed-off        power and peak maximum output power when amplifying radio        frequency signals;    -   an instantaneous bandwidth of at least 20 percent and an        efficiency of greater than 35 percent between 6 dB backed-off        power and peak maximum output power r when amplifying radio        frequency signals;    -   an instantaneous bandwidth of at least 20 percent and an        efficiency of greater than 40 percent between 6 dB backed-off        power and peak maximum output power when amplifying radio        frequency signals; and    -   an instantaneous bandwidth of at least 10 percent and an        efficiency of greater than 45 percent between 6 dB backed-off        power and peak maximum output power when amplifying radio        frequency signals.

Those skilled in the art will recognize improvements and modificationsto the embodiments of the present disclosure. All such improvements andmodifications are considered within the scope of the concepts disclosedherein and the claims that follow.

What is claimed:
 1. A Doherty amplifier comprising: a carrier splitteroutput and a peaking splitter output; a Doherty combining node; acarrier path comprising: carrier power amplifier circuitry; a firstplurality of lumped elements coupled between the carrier splitter outputand the carrier power amplifier circuitry; and a second plurality oflumped elements coupled with the carrier power amplifier circuitry andthe Doherty combining node; and a peaking path comprising: poweramplifier circuitry; a first plurality of lumped elements coupledbetween the peaking splitter output and the peaking power amplifiercircuitry; and a second plurality of lumped elements coupled with thepeaking power amplifier circuitry and the Doherty combining node.
 2. TheDoherty amplifier of claim 1, wherein the plurality of lumped elementsare configured to provide wideband functionality to the Dohertyamplifier.
 3. The Doherty amplifier of claim 2 wherein the Dohertyamplifier covers communication bands operating in a frequency bandbetween about 0.9 GHz to about 2.7 GHz.
 4. The Doherty amplifier ofclaim 2 wherein each of the plurality of lumped elements does notcomprise a transmission line.
 5. The Doherty amplifier of claim 1,wherein the plurality of lumped elements are configured to providemultiband functionality to the Doherty amplifier.
 6. The Dohertyamplifier of claim 1 wherein each of the plurality of lumped elementsare synthesized as a group to provide a desired performancecharacteristic.
 7. The Doherty amplifier of claim 1 wherein the carrierpath first plurality of lumped elements is configured to advance a phaseof the carrier signal and the peaking path first plurality of lumpedelements is configured to delay a phase of the peaking signal therebycausing the peaking signal to lag the carrier signal by approximately 90degrees when the carrier and peaking signals are respectively presentedto the carrier and peaking power amplifier circuitries and the carrierand peaking second plurality of lumped elements are configured torespectively impose compensated carrier and peaking phase offsetscausing the peaking and carrier signals to arrive at the Dohertycombining node for reactive combining to generate an output signal. 8.The Doherty amplifier of claim 7 wherein the respective compensatedcarrier and peaking phase offsets effectively reverse the approximately90 degree phase shift provided by the carrier and peaking firstplurality of lumped elements as well as remove carrier and peaking phaseoffsets, the carrier phase offset substantially corresponding to areactive component of an impedance presented to an output of the carrierpower amplifier circuitry at an intended operating frequency range orranges and the peaking phase offset substantially corresponding to areactive component of an impedance presented to an output of the peakingpower amplifier circuitry at the intended operating frequency range orranges.
 9. The Doherty amplifier of claim 8 wherein the impedancepresented to the output of the carrier power amplifier circuitry iseffectively a composite impedance provided by the carrier secondplurality of lumped elements, the peaking path, and a transformercoupled to the Doherty combining node at the intended operatingfrequency range or ranges and the impedance presented to the output ofthe peaking amplifier circuitry is effectively a composite impedanceprovided by the peaking second plurality of lumped elements, the carrierpath, and the transformer coupled to the Doherty combining node at theintended operating frequency range or ranges.
 10. The Doherty amplifierof claim 1 wherein the carrier path first plurality of lumped elementsadvances a phase of the carrier signal by approximately 45 degrees andthe peaking first plurality of lumped elements delays a phase of thepeaking signal by approximately 45 degrees such that the phase of thepeaking signal lags the phase of the carrier signal by approximately 90degrees when the carrier and peaking signals are respectively presentedto the carrier and peaking power amplifier circuitries.
 11. The Dohertyamplifier of claim 10 wherein the compensated carrier and peaking phaseoffsets effectively reverse the approximately 90 degree phase shiftprovided by the carrier and peaking first plurality of lumped elementsas well as remove the respective compensated carrier and peaking phaseoffsets, the carrier phase offset substantially corresponding to areactive component of an impedance presented to an output of the carrierpower amplifier circuitry at an intended operating frequency range orranges and the peaking phase offset substantially corresponding to areactive component of an impedance presented to an output of the peakingpower amplifier circuitry at the intended operating frequency range orranges.
 12. The Doherty amplifier of claim 1 wherein an instantaneousbandwidth of at least 15 percent and an efficiency of greater than 40percent between 6 dB backed-off power and peak maximum output power areprovided by the enhanced Doherty amplifier when amplifying radiofrequency signals.
 13. The Doherty amplifier of claim 1 wherein aninstantaneous bandwidth of at least 15 percent and an efficiency ofgreater than 40 percent between 6 dB backed-off power and peak maximumoutput power are provided by the enhanced Doherty amplifier whenamplifying radio frequency signals.
 14. The Doherty amplifier of claim 1wherein the first and second plurality of lumped elements of the carrierpath and the peaking path include capacitive elements and inductiveelements.
 15. The Doherty amplifier of claim 1, wherein each of thecarrier amplifier circuitries include gallium nitride.
 16. A multibandDoherty amplifier comprising: a carrier splitter output and a peakingsplitter output; a Doherty combining node; a carrier path comprising:carrier power amplifier circuitry; a first plurality of lumped elementscoupled between the carrier splitter output and the carrier poweramplifier circuitry; and a second plurality of lumped elements coupledwith the carrier power amplifier circuitry and the Doherty combiningnode; and a peaking path comprising: power amplifier circuitry; a firstplurality of lumped elements coupled between the peaking splitter outputand the peaking power amplifier circuitry; and a second plurality oflumped elements coupled with the peaking power amplifier circuitry andthe Doherty combining node; wherein each of the plurality of lumpedelements are configured to provide multiband functionality to themultiband Doherty amplifier.
 17. The multiband Doherty amplifier ofclaim 16 wherein each of the plurality of lumped elements does notcomprise a transmission line.
 18. The multiband Doherty amplifier ofclaim 16 wherein each of the plurality of lumped elements aresynthesized as a group to provide a desired performance characteristic.19. The multiband Doherty amplifier of claim 16 wherein the carrier pathfirst plurality of lumped elements is configured to advance a phase ofthe carrier signal and the peaking path first plurality of lumpedelements is configured to delay a phase of the peaking signal therebycausing the peaking signal to lag the carrier signal by approximately 90degrees when the carrier and peaking signals are respectively presentedto the carrier and peaking power amplifier circuitries and the carrierand peaking second plurality of lumped elements are configured torespectively impose compensated carrier and peaking phase offsetscausing the peaking and carrier signals to arrive at the Dohertycombining node for reactive combining to generate an output signal. 20.The multiband Doherty amplifier of claim 19 wherein the respectivecompensated carrier and peaking phase offsets effectively reverse theapproximately 90 degree phase shift provided by the carrier and peakingfirst plurality of lumped elements as well as remove carrier and peakingphase offsets, the carrier phase offset substantially corresponding to areactive component of an impedance presented to an output of the carrierpower amplifier circuitry at an intended operating frequency range orranges and the peaking phase offset substantially corresponding to areactive component of an impedance presented to an output of the peakingpower amplifier circuitry at the intended operating frequency range orranges.
 21. The multiband Doherty amplifier of claim 20 wherein theimpedance presented to the output of the carrier power amplifiercircuitry is effectively a composite impedance provided by the carriersecond plurality of lumped elements, the peaking path, and a transformercoupled to the Doherty combining node at the intended operatingfrequency range or ranges and the impedance presented to the output ofthe peaking amplifier circuitry is effectively a composite impedanceprovided by the peaking second plurality of lumped elements, the carrierpath, and the transformer coupled to the Doherty combining node at theintended operating frequency range or ranges.
 22. The multiband Dohertyamplifier of claim 16 wherein the carrier path first plurality of lumpedelements advances a phase of the carrier signal by approximately 45degrees and the peaking first plurality of lumped elements delays aphase of the peaking signal by approximately 45 degrees such that thephase of the peaking signal lags the phase of the carrier signal byapproximately 90 degrees when the carrier and peaking signals arerespectively presented to the carrier and peaking power amplifiercircuitries.
 23. The multiband Doherty amplifier of claim 22 wherein thecompensated carrier and peaking phase offsets effectively reverse theapproximately 90 degree phase shift provided by the carrier and peakingfirst plurality of lumped elements as well as remove the respectivecompensated carrier and peaking phase offsets, the carrier phase offsetsubstantially corresponding to a reactive component of an impedancepresented to an output of the carrier power amplifier circuitry at anintended operating frequency range or ranges and the peaking phaseoffset substantially corresponding to a reactive component of animpedance presented to an output of the peaking power amplifiercircuitry at the intended operating frequency range or ranges.
 24. Themultiband Doherty amplifier of claim 16 wherein the carrier path firstand second plurality of lumped elements and the peaking first and secondplurality of lumped elements are configured such that a frequencyresponse of the enhanced Doherty amplifier provides a first pass-bandfor a first communication band in a first frequency range and a secondpass-band for a second communication band in a second frequency range,which is separate from the first pass-band.
 25. The multiband Dohertyamplifier of claim 24 wherein an instantaneous bandwidth of at least 15percent and an efficiency of greater than 45 percent between 6 dBbacked-off power and peak maximum output power is provided by theenhanced Doherty amplifier when amplifying radio frequency signals inboth the first and second communication bands.
 26. The multiband Dohertyamplifier of claim 16 wherein an instantaneous bandwidth of at least 15percent and an efficiency of greater than 40 percent between 6 dBbacked-off power and peak maximum output power are provided by theenhanced Doherty amplifier when amplifying radio frequency signals. 27.The multiband Doherty amplifier of claim 16 wherein an instantaneousbandwidth of at least 15 percent and an efficiency of greater than 40percent between 6 dB backed-off power and peak maximum output power areprovided by the enhanced Doherty amplifier when amplifying radiofrequency signals.
 28. The multiband Doherty amplifier of claim 16wherein the first and second plurality of lumped elements of the carrierpath and the peaking path include capacitive elements and inductiveelements.
 29. The multiband Doherty amplifier of claim 16, wherein eachof the carrier amplifier circuitries include gallium nitride.
 30. Themultiband Doherty amplifier of claim 16 wherein the carrier path firstand second plurality of lumped elements and the peaking path first andsecond plurality of lumped elements are configured such that a frequencyresponse of the of the multiband Doherty amplifier provides a firstpass-band for a first communication band in a first frequency range anda second pass-band for a second communication band in a second frequencyrange, which is separate from the first pass-band.
 31. The enhancedDoherty amplifier of claim 30 wherein the frequency response provides adip between the first and second pass-bands.
 32. An enhanced Dohertyamplifier comprising: a plurality of enhanced Doherty modules whereineach of the enhanced Doherty modules is configured to receive an inputsignal and provide an output signal; a splitter output configured tosupply the input signal to each of the plurality of enhanced Dohertymodules a combiner configured to combine each output signal receivedfrom the plurality of enhanced Doherty modules; wherein each of theenhanced Doherty modules comprises: a carrier splitter output and apeaking splitter output; a Doherty combining node; a carrier pathcomprising a carrier power amplifier circuitry, a carrier input networkcoupled between the carrier splitter output and the carrier poweramplifier circuitry, and a carrier output network coupled between thecarrier power amplifier circuitry and the Doherty combining node; and apeaking path comprising peaking power amplifier circuitry, a peakinginput network coupled between the peaking splitter output and thepeaking power amplifier circuitry, and a peaking output network coupledbetween the peaking power amplifier circuitry and the Doherty combiningnode, wherein: the carrier and peaking input networks are configured toimpose phase shifts causing the peaking signal to lag the carrier signalby approximately 90 degrees when the carrier and peaking signals arerespectively presented to the carrier and peaking power amplifiercircuitries; and the carrier and peaking output networks are configuredto respectively impose compensated carrier and peaking phase offsetscausing the peaking and carrier signals to arrive at the Dohertycombining node for reactive combining to generate the output signal.